Capacitive measurement and/or detection systems have a wide range of applications, and are among others widely used for the detection of the presence and/or the position of conductive body in the vicinity of an electrode of the system. A capacitive sensor, called by some electric field sensor or proximity sensor, designates a sensor, which generates a signal responsive to the influence of what is being sensed (a person, a part of a person's body, a pet, an object, etc.) upon an electric field. A capacitive sensor generally comprises at least one antenna electrode, to which is applied an oscillating electric signal and which thereupon emits an electric field into a region of space proximate to the antenna electrode, while the sensor is operating. The sensor comprises at least one sensing electrode—which could comprise the one or more antenna electrodes themselves—at which the influence of an object or living being on the electric field is detected.
The technical paper entitled “Electric Field Sensing for Graphical Interfaces” by J. R. Smith, published in Computer Graphics I/O Devices, Issue May/June 1998, pp 54-60 describes the concept of electric field sensing as used for making non-contact three-dimensional position measurements, and more particularly for sensing the position of a human hand for purposes of providing three dimensional positional inputs to a computer. Within the general concept of capacitive sensing, the author distinguishes between distinct mechanisms he refers to as “loading mode”, “shunt mode”, and “transmit mode” which correspond to various possible electric current pathways. In the “loading mode”, an oscillating voltage signal is applied to a transmit electrode, which builds up an oscillating electric field to ground. The object to be sensed modifies the capacitance between the transmit electrode and ground. In the “shunt mode”, which is alternatively referred to as “coupling mode”, an oscillating voltage signal is applied to the transmit electrode, building up an electric field to a receive electrode, and the displacement current induced at the receive electrode is measured, whereby the displacement current may be modified by the body being sensed. In the “transmit mode”, the transmit electrode is put in contact with the user's body, which then becomes a transmitter relative to a receiver, either by direct electrical connection or via capacitive coupling.
The capacitive coupling is generally determined by applying an alternative voltage signal to a capacitive antenna electrode and by measuring the current flowing from said antenna electrode either towards ground (in the loading mode) or into the second electrode (receiving electrode) in case of the coupling mode. This current is usually measured by means of a transimpedance amplifier, which is connected to the sensing electrode and which converts a current flowing into said sensing electrode into a voltage, which is proportional to the current flowing into the electrode.
FIG. 1 shows a typical prior art circuit configured to measure an unknown capacitance in so-called ‘loading’ mode meaning that the capacitance between an electrode of a capacitive sensor and ground or earth is measured.
An AC voltage source 1 generates an AC voltage signal of known frequency and amplitude, for example a periodic sine wave of 100 kHz and 1 V peak amplitude. The output node 2 of AC voltage source 1 is connected to the non-inverting input of an operational amplifier 3. Operational amplifier 3 is configured as transimpedance amplifier. Operational amplifier 3, through the feedback action of associated feedback impedance 4 (preferably a capacitance connected in parallel to a resistance, whereby the impedance of the capacitance at the operating frequency is at least 10 times smaller than the resistance), maintains substantially the same potential on its inverting input as on its non-inverting input, thereby keeping sense node 5 at the same potential than AC voltage source output 2. The unknown capacitance 6 to be measured accordingly has the AC voltage source voltage applied across its “plates”.
The current flowing through unknown capacitance 6 is then given by its capacitance and the known AC voltage source voltage, said current flowing also through feedback impedance 4 as the input current into the non-inverting input of amplifier 3 is substantially zero.
The voltage on output 7 of amplifier 3 is accordingly responsive to the AC voltage source voltage and the unknown capacitance. This amplifier output voltage is then mixed with mixer 8 (for example a switching mixer or a multiplier) whereby the local oscillator input of mixer 8 is driven by the AC voltage source output 2. The output of mixer 8 is a DC voltage superimposed with multiples of the AC voltage source frequency, the DC voltage level being responsive to the amplitude of the amplifier output 7 and thereby of AC voltage source output voltage 2 and unknown capacitance 6.
As only the DC voltage is desired, the multiples of the AC voltage source frequency are filtered out with low pass filter 10. The output signal 11 of the low pass filter is a DC voltage responsive to the AC voltage source voltage and the unknown capacitance. Furthermore, an adjustable phase shift (preferably of selectable steps of 0 and 90 degrees) can be introduced between the AC voltage source output 2 and local oscillator input of mixer 8, thereby allowing the measurement of the complex impedance 6 instead of a capacitance 6.
FIG. 2 shows a typical prior art circuit configured to measure an unknown capacitance in so-called ‘coupling’ mode meaning that the capacitance between two electrodes of a capacitive sensor is measured.
In this variant, an AC voltage source 1 generates an AC voltage signal of known frequency and amplitude, for example a periodic sine wave of 100 kHz and 1 V peak amplitude. The output node 2 of AC voltage source 1 is connected to the first plate of unknown capacitance 6. The second plate of unknown capacitance 6 is connected to the inverting input of an operational amplifier 3. The non-inverting input of amplifier 3 is connected to ground. Operational amplifier 3, through the feedback action of the associated feedback impedance 4 (preferably a capacitance connected in parallel to a resistance, whereby the impedance of the capacitance at the operating frequency is at least 10 times smaller than the resistance), maintains substantially the same potential on its inverting input as on its non-inverting input, thereby keeping sense node 5 at ground potential. The unknown capacitance 6 to be measured accordingly has the AC voltage source voltage applied across its “plates”.
The current flowing through unknown capacitance 6 is then given by its capacitance and the known AC voltage source voltage, said current flowing also through feedback impedance 4 as the input current into the non-inverting input of amplifier 3 is substantially zero.
The voltage on output 7 of amplifier 3 is accordingly responsive to the AC voltage source voltage and the unknown capacitance. This amplifier output voltage is then mixed with mixer 8 (for example a switching mixer or a multiplier) whereby the local oscillator input of mixer 8 is driven by the AC voltage source output 2. The output of mixer 8 is a DC voltage superimposed with multiples of the AC voltage source frequency, the DC voltage level being responsive to the amplitude of the amplifier output 7 and thereby of AC voltage source output voltage 2 and unknown capacitance 6.
As only the DC voltage is desired, the multiples of the AC voltage source frequency are filtered out with low pass filter 10. The output signal 11 of the low pass filter is the DC voltage responsive to the AC voltage source voltage and the unknown capacitance. Furthermore, an adjustable phase shift (preferably of selectable steps of 0 and 90 degrees) can be introduced between the AC voltage source output 2 and local oscillator input of mixer 8, thereby allowing the measurement of the complex impedance 6 instead of a capacitance 6.
For both prior art circuits, the gain of the transimpedance amplifier formed by the operational amplifier 3 and the feedback impedance 4 is configured to be as large as possible in order to achieve low noise performance, and the DC gain of the signal chain stages following the mixer can subsequently be made comparatively low, to avoid DC offset problems. For example, in a practical implementation, for an operating frequency of 100 kHz and a source amplitude of 1 V, the feedback impedance would be chosen to be a capacitor of 100 pF in parallel with a resistance of 1 MΩ.
However, the output signal range of the operational amplifier 3 is limited, for example to an amplitude of 2 V peak for a 5 V power supply. This implies that a parasitic AC current injected into the sense electrode of the capacitive sensor of more than 126 μA peak amplitude will drive the operational amplifier into saturation and introduce an error into the measurement of the unknown capacitance. Such parasitic AC currents are e.g. generated by external noise sources, one example being the so-called ‘Bulk current injection’ (BCI) test during the qualification of an occupant detection system.